Tag Archives: supercapacitor

Supercaps For The Win!

A couple of years ago my parents bought themselves a LifeFitness R3 electronic exercise bike. It’s a pretty slick little machine, albeit expensive. The R3 has a plethora of workout and intensity options, a built-in heart-rate monitor, and displays that show distance traveled, calories burned, etc. What’s really cool is that all of the electronics are pedal-powered; who really wants to suck energy from the grid just to get a workout?

LifeFitness R3 Exercise BikeBut I’m not writing this post just to tell you about a neat exercise bike. I’m writing because I’ve figured out a way to make it even neater. Err, more neat. Whatever. Anyway, there’s one thing about the LifeFitness R3 that we’ve found slightly annoying: because it’s pedal-powered, whenever you stop riding, the display shuts off and the electronics reset. So if you’re in the middle of a 30-minute ride and get distracted by, let’s say, a Justin Bieber commercial, you’ll lose all of your stats and will have to re-start.

So since I hold degrees in electrical engineering, my parents asked me to make use of my education and remedy this inconvenience. My first step was to [carefully] crack open the display panel so that I could assess my options (scroll down for the full label legend):

LifeFitness R3 Circuit Board
It turns out this bike is controlled by our old friend the AVR ATMega128 (Label #1). This is the same chip found in the development board I used for my chronograph. It’s also very closely related to the AVR microcontroller used in the Arduino Mega. I tell you, it warms my heart to see this chip out in real products. Ah, but I digress…

Well having located the brains of the operation, I started looking for their power source. I quickly spotted two TO-220 package 5V linear regulators (labeled “R” in the image above). However, my multimeter (the only tool I had available at the time) indicated that neither of these were connected to the AVR. Eventually I located a tiny 8-SOIC regulator (Label #2) just beneath a pair of 2200uF power supply filter capacitors. A check of its pins indicated that this was in fact the device powering the AVR. And this regulator was fed by a set of wires that led down into the generator electronics. Interestingly, the generator appeared to be brushless, but I couldn’t get a good look at it or its electronics because of large panels I did not wish to damage (well, I thought about it).

I was also very excited to find a 2×5 ISP header on the main circuit board (Label #8). This meant that I might be able to reprogram the AVR to do my bidding. (Update: I’ve confirmed that this is possible; scroll down for details.) Perhaps I could have it enter power-save mode whenever the pedals stopped. Of course, this wouldn’t eliminate the power consumption of other devices on the board (display drivers, regulators, op-amps, etc). Plus, trying to reverse-engineer and modify machine code is no picnic (at least not to my knowledge). I decided to avoid this rabbit-hole and keep things simple.

NessCap 5V, 2.5F SupercapacitorsMy best option seemed to be the addition of supercapacitors. I could just tie them in parallel with the supply line filter caps. That way, the AVR’s regulator would continue to get stored power even after the user stopped biking. Adding capacitance to the regulator’s output was another option. However, the high initial charging current required by a large capacitor could be damaging to a device only rated to supply 100mA.

So I had two questions: how much capacitance do I need, and how much voltage will it have to handle? The second question was answered simply – I hooked my voltmeter up to the supply lines while pedaling and measured about 10.5VDC. To determine the amount of capacitance required, I used the following formula:

Ic = C*(dV/dT)

The ATMega128’s datasheet says it draws a current (Ic) of 19mA at 8Mhz and 5V, so let’s roughly double that figure just to be safe (to account for losses in the regulator and consumption by additional components). If the regulator can safely operate down to 5.5V, then our dV value will be 10.5 – 5.5 = 5V. Finally, let’s say we want to operate for 90 seconds. This means we need a capacitance of at least 0.72F. When I looked at Digi-Key (at the time), my best option was to purchase three 5V, 2.5F capacitors. Put in series, they’d be able to handle up to 15V, but their total capacitance would be reduced to 0.83F – still more than was necessary. Here’s a closeup image which shows the three supercaps linked together and soldered across one of the power supply filter capacitors:

Closeup of the supercapacitor fun-pack
So how did it all work? Splendidly. It turns out the AVR circuitry only drew about 30mA, giving approximately 120 seconds before full discharge. So now, whenever you hop off the bike to get water, adjust the stereo, or pet the dog, the bike’s display turns off (since it’s powered by a separate regulator), but the AVR continues to run, and will hold your current program and position for up to two minutes. A nifty feature added for about $20.

LifeFitness R3 Circuit Board
While I’m on the subject, I also found it interesting that the LifeFitness R3’s circuit board includes connections for a serial port (Label #9) as well as pins for a safety switch (Label #10). I suppose these were intended for other features not included with this model, but were left in place to reduce PCB manufacturing costs. For instance, the safety switch must have been meant for use with treadmills (I can’t see the need for this on a stationary bike). Perhaps the serial port is for a computer link of some sort? I’m tempted to test it out…

So finally, here’s the complete legend for the circuit board pictured above:

  1. ATMega128 microcontroller
  2. Linear regulator (8-SOIC) supplying the microcontroller
  3. Supercapacitor fun pack (3x 2.5F, 5V caps)
  4. Pushbutton circuit board
  5. Display driver IC (Holtek HT1647, 4-level grayscale, 64×16 LCD controller)
  6. Main I/O connector (includes power connections)
  7. Beeper (or, if you prefer, the annunciator)
  8. ISP header (for AVR programming)
  9. Serial port connections
  10. Safe switch connections

And because the quality of one’s post is directly related to the number of images contained therein, here’s a picture of my yellow lab. He’s not too sure about cameras just yet…

Marti (the Dog)

Update (10/8/2010): So I pulled out my old serial AVRISP with its 2×5 connector this afternoon, just to see if I could talk with the bike’s ATMega128. As it turns out, none of the chip’s lock bits were set, so I was able to download the HEX file with no problem (except for the strain on my arms while I kept the pedals turning). This means it is entirely possible for me to make modifications to the R3’s firmware. Of course, I’d have to figure out how to convert HEX back into ASM (which seems to be a questionable practice). If anyone else out there is interested in looking into this, feel free to leave a comment.

So You Want to Use PWM, Eh?

PWM Waveform Captured on an OscilloscopePulse-width modulation. It probably sounds a little confusing if you’re new to electronics. Kindof a word mashup, really. What do pulses, width, and modulation have to do with each other anyway? I remember first learning about PWM during my freshman year of college at RPI. I was in a pilot course called “Foundations of Engineering” under the excellent instruction of Professor Kevin Craig (whom I later worked for). I remember thinking later, “Hey, this PWM stuff is pretty clever!” So let’s take a look at PWM and see what we can learn. (If you’re already familiar with the basics of PWM, skip down a few paragraphs for more advanced topics and experiments!)

Say you’ve got a light-emitting diode (LED) and a battery. If you connect the two directly, the LED should produce a lot of light (assuming the voltage of the battery isn’t too high for the LED). But what if you wanted to reduce the amount of light that LED produces? Well, you could add a resistor in series with the LED to reduce the amount of current supplied by the battery. However, this won’t allow for easily adjustable brightness and may waste a bit of energy. That loss may not matter for a single LED, but what if you’re driving several high-power LEDs or light bulbs? This is where pulse-width modulation comes into play.

PWM Graph - 30% Duty CycleImagine you could connect and disconnect the LED and battery multiple times per second, causing the LED to flash or pulse (see graph above). If this ON-OFF cycle is fast enough, you won’t even notice the blinking. In fact, the LED will appear to be continuously lit, but reduced in brightness. In addition, its brightness will be proportional to the ratio of the on and off times. In other words, if the LED is connected for 30% of a pulse cycle, it will appear to be producing about 30% of its full brightness continuously, even though it’s actually turning completely on and off. So to adjust the brightness of the LED, all we need to do is adjust, or modulate, that ON-OFF ratio, also known as the pulse width – hence the name! The ratio between the on and off time is also commonly called the duty cycle.

Now in case you’re imagining yourself frantically flipping switches on and off, or tapping wires against battery terminals, you can stop. Just put a transistor in series with your LED! It can act as a switch which can be controlled by a microcontroller or some type of oscillator circuit (see links below).

Hobby Servo (Commanded via PWM)So what’s PWM good for, anyways? Well, dimming LEDs and other lights is just one of a number of applications (example). You’ll also find PWM used in motor controllers. You can make a very simple DC speed control using a PWM generator and a single transistor (examples – notice the extra diodes in use here to prevent damaging inductive spikes). In addition, PWM is very important for some types of power supplies; specifically the aptly-named “switched-mode” PSUs. This technique can also be used to create a digital to analog converter (DAC) by low-pass filtering the square wave. Finally, pulse-width modulation is sometimes used as a means of digital communication. For example, to command the position of a hobby servo.

Now you may be wondering why I’m writing about PWM all of a sudden. Well, there’s actually a point to all of this background information. By now, you’ve probably seen a car or two with these new-fangled LED tail lights. They’re pretty easy to spot since you can typically make out the individual LEDs within the whole tail light assembly:

Ford LED Tail Light Upgrade - Ain't that a Fancy Photo?
But have you ever noticed that on some cars (e.g. Cadillacs), these lights tend to flicker? You may not see it if you’re looking straight ahead, but if you quickly move your eyes from left to right, you may catch a glimpse of the flicker created by a low-frequency PWM controller. Now, call me strange, but I find this really annoying and distracting. Maybe I just have fast eyes or something, but I hate flicker. Back in the days of CRT monitors I could usually tell the difference between 60Hz and 70Hz refresh rates. But in the case of these tail lights, it sounds like there’s danger for people with photosensitive epilepsy. According to the Epilepsy Foundation, flashing lights in the 5 to 30Hz range can trigger seizures. Obviously, having a seizure while driving would not be a good thing for anyone.

By the way, if you’re ever trying to determine the frequency of a blinking light, just snap a couple pictures while moving your camera (or the light). The one catch is that you need to be able to specify a known shutter speed. Then you just have to count the blinks and divide by the shutter speed (in seconds) to find frequency. Here’s an example:

LED PWM Frequency Comparison

This method can also give you a pretty good indication of duty cycle – in this case it looks to be about 60%. Here’s a second shot I took while on the road one night. You can tell the streetlights are running on 60Hz AC (although they’re not LEDs so they never go completely dark during a cycle), while the green stoplight is likely getting DC:

Pulsing Streetlights

I’m thinking this long-exposure shot might also pass as modern art in some circles.

The Advanced Stuff

So what’s the deal with these awful low-frequency PWM tail lights? Well, one reason you might choose a lower frequency is to save on energy lost during switching. Both LEDs and the transistors used to drive them have parasitic capacitance. In other words, they store a very very small amount of energy (think nanojoules) each time you turn them on. This energy is consumed in addition to the steady-state power drawn by the LED to provide illumination. Furthermore, this stored energy is rapidly dissipated (and thus not recovered) each time the device turns off. Now if you’re turning an LED on and off fifty times per second, it’s probably no big deal. But what if you wanted to eliminate any possibility of flicker by driving the frequency up into the kilohertz range? Would this introduce substantial power loss? I was curious, so setup a simple experiment to find out.

Test Setup
The heart of this test circuit is fairly simple – two bright red LEDs (Model OVLBR4C7) along with 92Ω current-limiting resistors controlled by a BS170 MOSFET. To measure the power consumed by this circuit, I’ve taken a non-traditional approach. Because I was worried that the cheap ammeters I have available would be thrown off by varying PWM frequencies, I decided to measure power consumption based on the discharge time of a supercapacitor. And who doesn’t love supercaps, anyways?

The theory is pretty simple. The energy stored in a capacitor is equal to ½*C*V² (Joules). So all I had to do was charge up the cap, measure its voltage, let the circuit discharge it over a fixed period of time, then measure the final cap voltage. For my 2.5F capacitor (from NessCap), I chose ~60 seconds as my discharge period. Here’s a screenshot of the voltage logging application I used to collect my test data:

IOBoard Test Program
The white line in the graph above plots the capacitor voltage during discharge. The red line indicates the voltage measured across a phototransistor (L14C1). This was used to quantify the amount of light produced by the LEDs at each test point. To get a better measurement I covered the LEDs and phototransistor with an opaque plastic cup, then covered the whole setup with a shoebox and turned off the lights. I was trying to see if, for some reason, the intensity of the LEDs was non-linear with respect to duty cycle or was affected by PWM frequency. Unfortunately this data turned out to be rather boring, but I’ve still included it in my summary spreadsheet which you can download below.

Now before I go on, you’re probably wondering what sort of data acquisition hardware I’m using. Well I doubt you’ve heard of it as it hasn’t yet been commercially released. Right now it’s being called the RPI IOboard. It’s a pretty impressive piece of hardware with dual 12-bit, 1.5MSPS ADCs, dual 14-bit, 1.4MSPS DACs, and a host of digital I/O all powered by a 400Mhz Blackfin processor. For the past few years it’s been developed at RPI and tested at a number of schools across the country. However since the project’s lead professor, Don Millard, left RPI last year, I’m not exactly sure what will become of the board. The screenshot you see above is actually one of several executable VIs I developed as examples for use with the board. Further information on the hardware can be found here.

Test Setup Closeup
So back to the experiment at hand. For my first round of testing, I utilized the IOBoard to generate varying PWM signals for the MOSFET. Thus, the current required to drive the BS170 was not included in my first measurements. I varied both frequency and duty cycle for three pairs of LEDs: white (C513A-WSN), red (OVLBG4C7), and green (OVLBR4C7).

TABLE 1: Data for power consumption tests without gate-drive losses:

Frequency/Duty Cycle (WHITE LED) 30% 60% 90%
50 Hz
36.15 mW 62.08 mW 84.89 mW
300 Hz
36.26 mW 63.50 mW 85.12 mW
10 kHz
38.75 mW 64.25 mW 86.14 mW
100 kHz
38.52 mW 62.80 mW 86.59 mW
Frequency/Duty Cycle (RED LED) 30% 60% 90%
50 Hz
54.70 mW 93.82 mW 123.75 mW
300 Hz
57.76 mW 93.81 mW 125.35 mW
10 kHz
56.99 mW 94.00 mW 126.08 mW
100 kHz
56.61 mW 95.11 mW 125.47 mW
Frequency/Duty Cycle (GREEN LED) 30% 60% 90%
50 Hz
41.49 mW 71.29 mW 91.65 mW
300 Hz
41.93 mW 70.29 mW 91.69 mW
10 kHz
41.90 mW 69.96 mW 93.36 mW
100 kHz
42.57 mW 69.71 mW 93.58 mW

So if you look through the data above, you’ll notice that there is, on average, a slight positive correlation between power consumption and frequency. In other words, the higher the switching frequency, the greater the power consumption. This is just what we would expect. Again, this data does not include losses due to transistor gate capacitance, only losses due to the LEDs’ capacitance and the MOSFET’s output capacitance.

For my next test, I wanted to see what losses might be incurred in driving the MOSFET’s gate. Thus, I called on my trusted 8-bit AVR microcontroller (ATMega644P). I wrote a very simple program (which may be downloaded below) to produce a varying PWM output from one of the MCU’s timer/counter outputs. I then measured the power consumption of the entire circuit, AVR included. For this test I only used a 60% duty cycle:

TABLE 2: Data for the ATMega644 driving a BS170 and two green LEDs:

Test Frequency Total Average Power (mW) Calculated Switching
Losses (mW)
50 Hz
91.741 0.000
300 Hz
92.708 0.000
10 kHz
92.622 0.016
100 kHz
92.978 0.157
1 Mhz 95.789 1.568

TABLE 3: Data for the ATMega644 driving a FDP8860 and two green LEDs:

Test Frequency Total Average Power (mW) Calculated Switching
Losses (mW)
50 Hz
93.475 0.004
300 Hz
95.809 0.021
10 kHz
98.238 0.710
100 kHz
114.526 6.848
1 Mhz 161.657 60.914

TABLE 4: Data for the ATMega644 directly driving two green LEDs:

Test Frequency Total Average Power (mW) Calculated Switching
Losses (mW)
50 Hz
69.278 0.000
300 Hz
67.926 0.000
10 kHz
68.778 0.015
100 kHz
68.534 0.147
1 Mhz 70.708 1.467

Discussion of Results

In Tables 2-4, we’re starting to see a much clearer positive correlation between frequency and power consumption. For these tests I also added a fifth data point not gathered with the IOBoard: a frequency of 1Mhz. This should in theory increase our maximum losses by 10x. The results seem to support with this prediction.

The tables above also include a rudimentary calculation for switching losses based on capacitances. I measured the capacitance of my green LEDs to be about 120pF (this value was not mentioned in the datasheet). The gate capacitance of the BS170 is given in its datasheet as 24pF. Finally, the input capacitance of the FDP8860 (a much beefier power MOSFET) is typically listed as 9200pF. To determine switching losses I again applied the formula for a capacitor’s stored energy (½*C*V²). At each switching interval, the parasitic capacitances in the circuit store and then dissipate this much energy. So to determine how much power is lost, we simply multiply this lost energy by the switching frequency (since 1 watt = 1 joule/sec). It appears that these calculated figures match the measurements fairly well. Isn’t it nice when math agrees with reality? Gives me a fuzzy feeling, that.

Now we can essentially think of the 50Hz test point as a baseline with zero switching loss. For the data in Table 4, the 50Hz power consumption is about 69.3mW. The calculation predicts that at 1Mhz, we’ll lose 1.5mW to parasitic capacitance for a total consumption of 69.3 + 1.5 = 70.8mW. This isn’t that far from our measured 70.7mW.

It’s also interesting to note the substantially higher losses incurred when using the FDP8860. This is largely due to its (relatively) enormous input capacitance of 9200pF. This is nearly 400x the capacitance of the tiny BS170. That’s the price you pay for the ability to sustain larger currents without overheating. For more information on power MOSFETs have a look at this IRF document called “Power MOSFET Basics.”

Summary

Well after all that, I’m going to say that whoever manufactures these tail lights can’t really use efficiency as an excuse for choosing a low switching frequency. Unless they need huge FETs to drive huge currents, switching losses really aren’t so much of an issue. I’m guessing that somehow it was just cheaper to go with a low frequency. I’m pretty sure the components themselves aren’t any cheaper, but perhaps the assembly was less expensive. It may be that some automakers already had a low-frequency module in place to drive old incandescent bulbs and then when LEDs came along they just kept using that same module. Anybody out there care to comment on this?

So my advice to those making LED dimmers: pick a frequency of about 300-500Hz to eliminate flicker while keeping switching loss low. Then find yourself a sufficiently large transistor with low capacitance and low on-resistance. And if you’re working on motor controls or power supplies, things get a lot more interesting, but as a start, try a frequency in the 20+ kHz range to avoid audible whine. Good luck!

  • For further reading on LED losses, try this NI article: Light Emitting Diodes.
  • For more accurate MOSFET swithing loss formulae, try this MAXIM article.
  • Test code for the ATMega644P is available here.
  • A complete spreadsheet containing all data can be downloaded here.

Update (9/22/2010): In the comments below, Jas Strong pointed out that in my switching loss calculations, I’d also neglected the power lost in the MOSFET during turn-on. Jas is absolutely correct about that; I should have mentioned this previously. Essentially, while the gate capacitance of the MOSFET is charging, the resistance between drain and source will pass from very high to very low resistance as the conduction channel is formed. This time period, although short, includes a region of, shall we say, “moderate” resistance which briefly dissipates additional power.

Now, in the case of my two-LED test setup, I neglected the effects of resistive switching loss because they’re quite small. Let’s take a quick look at the numbers. First, we need to know how long it takes Vgs to reach the threshold voltage. For simplicity, I’m going to assume that my AVR drives the gate with a constant current of 40mA (the maximum an AVR will provide per I/O pin). Our worst-case turn-on time will occur with the FDP8860, which has a gate capacitance of 9200pF and a typical threshold voltage of 1.6V. Using the formula ic = C*(dv/dt), I find dv/dt = 4,347,826 which means we reach Vth in 1.6/4,347,826 = 368ns. At a switching frequency of 1Mhz, this represents about 37% of a switching cycle. However, we need to double this since we lose power durning turn-on and turn-off. Thus, we’re losing energy in the MOSFET’s resistance over 74% of a single cycle at 1Mhz. That sounds like a lot, but just how much energy is actually lost?

To determine this loss, I’m going to make a big assumption and say that the MOSFET ramps linearly from 20kΩ down to 0Ω during turn-on. I’m also going to assume the voltage of the diode is constant at 3V and the power supply is constant at 4.2V. Remembering that I have 92Ω resistors in series with the LEDs, the instantaneous power dissipation in the FET becomes 2*Rmos*[(4.2-3)/(92+Rmos)]^2 (based on the fact that I have two LEDs and using the formula P = RI^2 and ohms law, I = V/R). Now I need to integrate to determine an average power dissipation over this interval. If my math is correct (feel free to check me), I get a loss of 0.632mW. Since this occurs during 74% of a cycle, the total loss at 1Mhz will be about 0.468mW. Not too serious in my opinion.

Now of course, the power required by my two-LED setup is piddly in comparison with that drawn by a couple brake lights. Once you start sinking more current into your LEDs, this resistive switching loss, as well as the on-resistance of your MOSFET, is going to start to make a bigger difference. So thanks very much Jas for pointing this out!

Frequency Duty Cycle Start Cap Voltage Start Phototransistor Voltage
50 0.3 4.248407 1.464428967
300 0.3 4.246836767 1.4911225
10000 0.3 4.2389857 1.4911225
100000 0.3 4.243696367 1.538228733